Animal Location Tracker (A3038)

© 2020 Kevan Hashemi, Open Source Instruments Inc.

Contents

Description
Design
Set-Up
Current Consumption
Modifications
Development

Description

The Animal Location Tracker (ALT, A3038) provides an array of fifteen detector coils that receive signals from our subcutaneous transmitters and implantable stimulators. Above the coil array is a platform upon which we place a cage containing one or more animals with implanted transmitters. The A3038 decodes and records the transmitters signals and measures the radio-frequency power received by each of its fifteen detector coils, so as to obtain not only the signal value, but also an estimate of where the transmitter is located above the tracker platform. The A3038 is designed to measure the activity of animals in cages, and to make it possible to identify animals in video by correlating their movements with the movements of their implanted transmitters. Each transmitter has its own unique channel number, so each animal has its own unique location and velocity.


Figure: Prototype Power Detector and Demodulator (A3038X).

Our implanted transmitters emit 7-μs bursts of electromagnetic radiation in the range 902-928 MHz. Each burst contains a digital message. Each detector coil provides a measurement of radio-frequency power in this same frequency range. When one of the detector coils reports that a message transmission is in progress, all fifteen detector coils record the power they are receiving. The A3038 saves the transmitted message, along with fifteen eight-bit power values, in its memory, available for download by the Recorder Instrument in the LWDAQ Software, or download and storage to disk by the Neuroarchiver Tool. The Neuroarchiver contains a Tracker button that opens the Neurotracker Panel, which displays the measured position of transmitters on the tracker platform.


Figure: Neurotracker Panel Provided by the Neuroarchiver. Default parameters to control the location calculation are shown along the top.

The A3038 is an improved version of our original A3032 ALT. The A3032 provided fifteen detector coils on a 32 cm × 16 cm grid, and used the coils to measure transmitter position, but did not itself decode the radio-frequency transmissions. The A3038 uses each detector coil both as a power meter and a data receiver. We present the basis of the ALT position measurement in the A3032 Feasability Study and Development pages. We concluded that absolute accuracy of the ALT's position measurement for a transmitter in a beaker of water with respect to the coordinate system defined by its coil centers is ±25% of the coil pitch 90% of the time, and ±100% of the coil pitch the rest of the time. Because there is no way to know whether the current absolute position is accurate or not, the absolute position measurement is not useful. Its measurement of the direction and magnitude of movement, however, provide a reliable measurement of animal activity, and a reliable way to identify animals in video. The correlation between the ALT measurement of movement and video blob-tracking permits us to be 100% certain which blob corresponds to which animal, even when there are a dozen animals in the field of view.

Version X (cm) Y (cm) Coil Pitch (cm) Coil Type Num Coils Comment
A3038A 50 25 12 33 nH SMT 15 100.0% accurate disambiguation for height 0-5 cm.
Table: Versions of the A3038.

The A3038 communicates with our data acquisition computer through an ethernet connection that also provides power for its detectors, amplifiers, and data buffer. We use a standard Power Over Ethernet (PoE) switch to provide communication and power delivery for multiple A3038s installed in faraday enclosures. We connect our data acquisition computer to the same switch, and so download the signals and power measurements from multiple A3038s.

Design

S3038A_1: RF Power Detector and Receiver for A3038A.
S3038A_2: Logic and Communications for A3038A.
S3038X_1: Prototype power detector and demodulator.
A303801X.zip: Gerber Files for A3038X printed circuit board.
A303801X_Top: Top view of A303801X printed circuit board.
A303801X_Bottom: Bottom view of A303801X printed circuit board.
LT5534: Radio-frequency power detector.
ADC081S101: Eight-bit serial ADC.
BGA2803: Low-power 23-dB gain DC-2GHz amplifier.
2014VS: Vertical, surface-mount inductor.

Set-Up

Current Consumption

[08-SEP-20] Current consumption fo the A3038X detector and demodulator is 32 mA for 3.0-V supply voltage.

Modifications

[19-SEP-20] The A3038X requires the addition of four 10 nF and four 47 pF decoupling capacitors for U1-U4, as well as two 100 pF capacitors around U6.

Development

[14-JUL-20] We examine the response of the A3032C ALT amplifier and detector. We apply a −6 dBm sweep 840-980 MHz to a Loop Antenna (A3015C). We generate the sweep with a Modulating Transmitter (A3014MT), and we split the sweep and mix with 910 MHz to produce an IF ±70 MHz, which we run through a 21-MHz low-pass filter before viewing on the scope. We hold the loop antenna above Coil 10 on V0384 and observer the following response on PW, which is U1004-3.


Figure: Sweep Response of Power Measurement. Yellow: PW on U1004-3, 500 mV/div. Green: Ramp voltage that controls A3014MT. Blue: IF reference, center is 910 MHz, left and right edges of bulge are 890 MHz and 930 MHz respectively.

We repeat on all fifteen coils and find the same response on each one, ±100 mV variation in 890-930 MHz, which is roughly ±3 dB. We remove the A3051C loop antenna and replace it with a 3-dB attenuator and a 50-mm bent wire antenna. We observe the same ±3 dB variation on power through the pass band. But in rare orientations of the antenna, all surrounding objects remaining stationary, received power drops suddenly, and variation is ±6 dB.

[29-JUL-20] We are considering using detector diode such as the SMS7630 in our coil amplifier to provide power limiting and power detection.


Figure: Behavior of the SMS7630-061 Schottky Diode. Left: Current versus Forward Voltate. Right: Rectified Voltage versus Input Power.

In SkyWorks application note APN1014, we see the detector circuit they used to obtain the above rectified voltage versus input power. They deliver power from a 50-Ω source, but do not load the source with 50 Ω. Instead, they place a diode and balast capacitor in series with the 50-Ω source impedance. The voltage across the diode will be roughly double what we would see if we loaded the source with 50 Ω. Their "Incident Power (dBm)" is the power that reflects off the detector circuit, which would be equal to the power delivered to a 50 Ω load. The "video resistance" they refer to in the detector plot is the resistor loading the balast capacitor.

We assemble a power detector made out of an SMS7630 diode and a 100 pF balast capacitor attached to a 50-Ω transmission line carrying the output of an A3029B amplifier. We measure the voltage across the balast capacitor, which we call the rectified voltage. We vary the transmission line power from −30 dBm to +28 dBm. Below −30 dBm our rectified voltage is swamped by noise. At +28 dBm, our amplifier is saturating. When we remove the diode and capacitor, our amplifier saturates at +30 dBm. With +6 dBm we add 1 kΩ in series with the diode and see only 90 mV, compared to 360 mV with no resistor.


Figure: Rectified Voltage versus Input Power for SMS7630 Power Detector. Incident power is terminated by a resistor. In parallel with the resister is the diode and a capacitor.

We solder 51 Ω from the center pin of a BNC socket to ground. In parallel we place a SMS7630 diode and a 1.0 nF balast capacitor. We supply power down a 1-m coaxial cable from our synthesizer, and vary power from −30 dBm to +10 dBm. We measure the voltage rectified voltage versus power and plot.

[30-JUL-20] We solder a 33 kΩ resistor across a BNC socket. In parallel we place a SMS7630 diode and a 1.0 nF balast capacitor. We connect the detector directly to the output of our synthesizer. Without the 33 kΩ, we see no sustained rectified voltage, because the incoming power is capacitively coupled.

[07-AUG-20] We have the schematic of a prototype detector coil circuit, which provides both power measurement and demodulation of SCT messages, S3038X_1. Top view of circuit board here. This circuit will provide the D input for an A3007D so as to provide SCT signal reception, and will also produce an output P from the power meter. Power supply will come from a two-pin molex plug.

[04-SEP-20] We assemble a prototype power detector and demodulator (A3038X), as in S3038X, except we omit U5. We remove L1 so that we can supply A through J1. We remove R9 so that we extract B through J2. We connect 3.0V to P1. We see 33 mA flowing in. With four of BGA2803 and one of LT5534 we expect 31 mA. We connect J2 to our hand-held spectrum analyzer, and J1 to our frequency synthesizer, with attenuators as needed, to obtain the following plot of B versus A for 910 MHz.


Figure: B versus A at 910 MHz.

The BGA2803 gain at 910 MHz is 24 dB. We have three 3-dB attenuators for −9 dB. We have the insertion loss of the SAQ filter L2, a B3588, which is around 2 dB. We expect the gain of the first two stages to be 37 dB. We see 41 dB from −90 to −50 dBm input. The output of U2 saturates at −5 dBm, which is consistent with the BG2803 data sheet.

We remove R1. Now we are shorting the input to ground with R17. We see 0.5 V at the output of the power detector, P, indicating −49 dBm at the input of the power detector, which implies −90 dBm at the input of U1, which is consistent with 50-Ω thermal noise in 900-930 MHz. When we restore R1, with our frequency synthesizer still attached, we see 1.8 V on P, meaning B is −17 dBm, which suggests interference of −58 dBm.


Figure: Response of LT5534 Power Detector.

We apply −60 dBm to A and measure B as we vary frequency. We see the pass-band of the SAW filter clearly. We are surprised by the higher gain at 900 MHz and 930 MHz, which mark the edges of the pass-band. We restore R2 and sweep frequency again, this time measuring power at J4.


Figure: Power versus Frequency for −60 dBm Applied to A. Orange: power at B with R9 removed. Green: power at J4 with all resistors loaded.

Output J4 is flat to within ±1 dB in the SAW filter pass band, at around −6 dBm. Diode arrays DA1 and DA2 are SMS7630. Their saturation current is 5 μA, so we expect their dynamic resistance to be 50 Ω for forward voltage 110 mV. Our guess was they would limit the amplitude to −3 dBm. But B and J4 are limited to −5 dBm by the BGA2803 saturation alone. We predict that removing DA1 and DA2 will change nothing. We would like B to lie in the range −60 dBm to 0 dBm when detecting transmitter power bursts. But we can receive up to −20 dBm from a transmitter held close to a detector coil. The gain from A to B is 21 dB too high.

[07-SEP-20] We apply −52 dBm 910 MHz to A. We have L1 and R9 removed. We measure −10 dBm at B. Gain is 42 dBm. We mix with +7dBm of 880 MHz using a ZAD-11 and see IF amplitude is 50 mVrms, or −13 dBm (conversion loss only −5 dBm). We sweep the frequency from 820-980 MHz and obtain the following IF trace.


Figure: Gain vs. Frequency for Path A to B. We apply −52 dBm sweep 820-980 MHz to A and mix B with 980 MHz to produce IF frequency (Red) and detector output P (Blue). The sweep is controlled by a ramp (Yellow). Screen center is 915 MHz.

Gain in this measurement is constant to ±2 dB in the pass-band of the SAW filter. We do not see the +10 dB ears that appear in our measurement using our hand-held spectrometer. For −10 dBm we expect 2.2 V at P. We see 2.4 V. We vary power supply voltage and measure gain and supply current.


Figure: Gain (dB) and Current Consumption (mA) with Power Supply Voltage (V). We do not have U5 loaded, but its consumption adds only 1 mA.

We restore R9 and apply −36 dB sweep 820-980 MHz to A. We mix the same sweep with 910 MHz to produce IF that we pass through 21 MHz low-pass filter. We have L3 = 3.3 nH, C16 = 0.5 pF, C17 = 2.0 pF, C18 = 10 pF, R25 omitted. We have U5 loaded, so that D = 4R. We look see the following tuning curve on D.


Figure: Demodulated (Green, D) and Power (Blue, P) vs. Frequency. We apply −36 dBm sweep 820-980 MHz to A. Red trace is sweep mixed with 915 MHz and low-pass filtered to ±21 MHz. Screen center is 915 MHz. We have C18 = 10 pF and R25 omitted.

We see D go from 0.5 V at 894 MHz to 2.5 V at 936 MHz, which is perfect for demodulating SCT signals. We apply an SCT signal −31 dB, to A. With no R25 and C18 = 10 pF, our SCT demodulated signal is a triangle wave. The rise time of D appears to be around 200 ns. Add R25 = 1 kΩ and see the trace below, rise time around 50 ns.


Figure: Demodulated SCT Signal (Green, D) and Power (Blue, P) vs. Time During SCT Transmission. We apply −31 dBm SCT signal centered on 914 MHz. Yellow: SCT bit trigger. We have C18 = 10 pF and R25 = 1 kΩ.

The above signal is similar to the traces we see in our downshifting receivers. We repeat our frequency sweep to see again the tuning response. We now have D going from 0.2 to 0.9 V in the band 894-936 MHz.


Figure: Demodulated (Green, D) and Power (Blue, P) vs. Frequency. We apply −36 dBm sweep 820-980 MHz to A. Red trace is sweep mixed with 915 MHz and low-pass filtered to ±21 MHz. Screen center is 915 MHz. We have C18 = 10 pF and R25 = 1 kΩ.

We try C18 = 0 pF with R25 omitted. We see the same demodulation sweep amplitude as for C18 = 10 pF with R25 omitted, but when we apply an SCT signal, the rise time is around 100 ns.

[10-SEP-20] We remove R1. We connect C13 to R6/R7 instead of B. We see 0.74 V on P. We load 100 pF from U6-4 to 0V and see 0.40 V. Adding another 100 pF makes no difference, nor does adding 10 pF. We load 100 pF across P1 and see 0.35 V on P. As soon as we restore R1, even without L1, we see 1.1 V. If we connect 50 Ω to J1 we see 0.44 V. With L1 loaded and J1 open circuit 1.1 V. Restore C13 to its previous connection and see 2.1 V on P. Remove R1 and see 1.2 V.

[12-SEP-20] Restore R1 and see 2.1 V on P. We test the hypothesis that J1 and L1 are picking up radio waves transmitted by J4, and the 80-dB total gain of the amplifier is generating oscillations at a frequency above 2.4 GHz. We rotate C7 and C10 so that they connect U3-6 and U4-6 to 0V. We still see 2.1 V on P. We rotate C4 to connect U2-6 to 0V. We connect C13 to R7, so we have U2, U3, U4 with grounded inputs, their outputs loaded by attenuators, and U6-6 driven by R7. With L1 loaded and J1 open circuit, P = 1.1 V. Remove L1, P = 1.1 V. J1 is not picking up radio waves from J4.

[14-SEP-20] We restore R1, restore C13 to B, and restore C4. Now we have amplifiers U1 and U2 working, driving R9/R10 attenuator, and also driving our spectrometer through J2. Amplifiers U3 and U4 have their inputs grounded with capacitors. The A3038X and spectrometer are inside a Faraday enclosure, and we have a 10-dB attentuator between J2 and the spectrometer. We see P = 2.1 V and B has peaks 1053 MHz −10 dBm, 886 MHz −50 dBm, and 929 MHz −55 dBm. We replace L2 with a wire link. We see P = 1.9 V. The spectrum of B contains peaks 1226 MHz −11 dBm, 1297 MHz −22 dBm and several others. Remove R1 and see P = 1.6 V and B has peaks 1220 MHz −21 dBm, 1261 MHz −26 dBm. Restore L2 and remove J1. See P = 2.1 V and B has peak 1061 MHz −4 dBm. Remove U1 and replace with wire link, P = 0.8 V, B has peak 1217 MHz −42 dBm. Replace L1, P = 2.1 V, B has peak 1037 MHz −7 dBm. Load 27 Ω for R1, R2, and R17. Now P = 0.8 V, B has peak 1221 MHz −42 dBm. Connect test transmitter, transmit off, peak 929 MHz −44 dBm. Turn on −38 dBm transmit signal and see P rising to 2.2 V. We load R1 = R2 = 15 Ω and R17 = 75 Ω. With J1 open circuit, B has peak 1010 MHz −10 dBm. With J1 connected to 50-Ω coax peak is 929 MHz −52 dBm and B is 0.8 V. With L1 loaded and J1 open, P = 1.8 V and B has peak 961 MHz −17 dBm. We load UPC2746T in place of U1, and restore R1, R2, and R17. With J1 open and L1 loaded, B has peak 840 MHz −12 dBm. Connect J1 and peak vanishes. Restore BGA2803 for U1 and load 0 Ω for R1/R2 and 50 Ω for R17. On P 2.1 V, on B 1012 MHz −8 dBm. Remove R1 and see P = 0.9 V. Remove R2, P = 0.9 V. Load R1 = R2 = R17 = 51 Ω. Have P = 1.0 V, B peak 929 MHz −46 dBm. Connect −68 dBm SCT signal and see P rise to 1.2 V during burst.

With L1 and J1 loaded, and R1 = R2 = R17 = 51 Ω, apply −8 dBm SCT signal to A3015C loop antenna. A3038X in Faraday enclosure. Can see signal clearly on B at range 50 cm. Take A3038X out of enclosure and hook up to P and D as well as trigger from SCT. Apply −48 dBm SCT to J1. See SCT signal clearly on DA, and P rises from interference level 1.5 V to burst level of 2.0 V. Disconnect SCT and connect spectromter to B, see −27 dBm in 902-928 MHz, with P varying 1.0-1.5 V. Connect −8 dBm SCT signal to A3015C. Reception range outside Faraday enclosure is only 10 cm.

[16-SEP-20] We restore R1 = R2 = 8.2 Ω and R17 = 150 Ω. We remove L1. We see P = 2.22 V as the circuit oscillates. We have 47 pF capacitors in P0603 package (Digikey 445-1277-1) with self-resonant frequency 900 MHz. We connect directly from pin U1-2 to C3-2 to add local decoupling. Now P = 1.24 V. We add 47 pF to U2 in the same way, P = 1.16 V. We load a second 47 pF in parallel to the one next to U1 and P increases to 1.64 V. So we remove that capacitor and add one to U3 and U4, so each has its own local 47 pF. Now P = 1.10 V. Again we double-up 47 pF next to U1, again P increases to 1.64 V. We move the two 47 pF and mount them on C3, P = 2.02 V. Remove one 47 pF, leaving one 47 pF on C3, P = 2.08 V. Replace C3 = 100 pF and the 47 pF addition with a single 47 pF, P = 2.04 V. Restore 47 pF next to U1, and load 47 pF for C3, see P = 1.84 V, which turns out to be −21 dBm of 1200 MHz. Replace C3 with 1.0 nF, P = 1.16 V. We remove R1 and P = 1.02 V. We replace C3, C5, C8, and C11 with 1.0 nF and each of U1-U4 has 47 pF soldered to pin two. We see P = 1.04 V. Now we double up the 47 pF on U1, and still see P = 1.04 V. So we remove the two 47 pF and stil see P = 1.04 V. Remove all 47 pF and P = 1.12 V. We restore the 47 pF and see 1.04 V again.

We have R1 = R2 = 8.2 Ω and R17 = 150 Ω, no L1, and combined 47 pF and 1.0 nF decoupling capacitors. We connect −60 dBm of 910 MHz through a 1-m cable to J1 and see −20 dBm at J2, suggesting gain at least 40 dB. At J3 we see +0 dBm and at J4 +2 dBm. Apply −40 dBm input and see +0 dBm at J3 and +1 dBm at J4. Outputs J3 and J4 are after −3 dB attenuators, and we are adding an additional 50-Ω load when we make our measurement, so saturated output power of U3 and U4 appear to be +3 dBm and +4 dBm. According to the BGA2803 data sheet, saturated output power should be −3 dBm. We apply SCT signal to J1 and find we need at least −38 dBm to get demodulated signal at D, even in Faraday enclosure, even when waiting for interference to subside.

[17-SEP-20] We go back to decoupling with 100 pF on their footprints and restore R1 = R2 = R17 = 51 Ω. We apply −58 dBm SCT signal to J1 and see clear demodulated levels on D. We apply −60 dBm from our synthisizer and measure power at B, J3, and J4 by plugging our spectrometer into J2, J3, and J4. We do not remove any resistors. When we plug our cable into J2, J3, or J4, the signal at the connector is loaded by two 50-Ω impedances in parallel. We measure power at each frequency by finding the peak in our spectrometer and observe the following ±5 dB variation in gain in the SAW filter pass-band. (NOTE: On 24-SEP-20 we obtain ±2 dBm gain uniformity within the SAW passband after exchanging L2.)


Figure: Power versus Frequency at Various Points. We have −60 dBm being delivered to A through J1, R1 = R2 = R17 = 50 Ω, no L1, all other resistors loaded. (NOTE: On 24-SEP-20 we obtain ±2 dBm gain uniformity within the SAW passband after exchanging L2.)

[18-SEP-20] We have R1 = R2 = R17 = 51 Ω, and no L1. Now we load in parallel with the C3, C5, C8, and C11 10 nF and 47 pF, with the 47 pF right next to the amplifier pins. We have no L1. With J1 open-circuit, P = 0.58 V. We detect −58 dBm SCT clearly on D when 929 MHz interference we pick up with spectrometer antenna is −60 dBm. With −47 dBm interference, D is contaminated with what looks like 50 MHz noise. We apply −47 dBm sweep to J1 and observe D and P.


Figure: Demodulator (Green) and Detector (Blue) Output for −46 dBm Sweep Input. We apply sweep to A, L1 not loaded. Red trace is sweep mixed with 915 MHz and low-pass filtered to ±21 MHz.

When we drop the sweep to −56 dBm, we must wait until 929 MHz interference dies down to −69 dBm before taking our photograph.


Figure: Demodulator (Green) and Detector (Blue) Output for −56 dBm Sweep Input. We apply sweep to A, L1 not loaded. Red trace is sweep mixed with 915 MHz and low-pass filtered to ±21 MHz. (NOTE: On 24-SEP-20 we obtain ±2 dBm gain uniformity within the SAW passband after exchanging L2 and using 820 MHz LO to observe B.)

We load L1 and place A3038X in Faraday enclosure. We look at D while moving a transmitter from one place to another. With the door closed, we see the transmit burst in all locations, and the data bits are clear for ranges up to 10 cm. When the transmitter is farther away, the bits are overwhelmed by 50 MHz noise. We note that the A3027E's superhet receiver provides a total gain of 100 dB with limiting at 6 dBm (±0.7 V) before demodulation, while this amplifier provides 80 dB gain before limiting at −5 dBm at J4.

[21-SEP-20] We have R1 = R2 = R17 = 51 Ω, L1 loaded, apply −40 dBm of 915 MHz to J1. See P = 2.12 V. Load 0.5 pF parallel with L1, see 2.12 V. Load 1.0 pF see 1.40 V, load 1.5 pF see 1.32 V. Remove parallel capacitance.

We have R1 = R2 = 8.2 Ω, R17 = 150 Ω, L1 loaded. Apply −40 dBm 915 MHz, P = 2.36 V (−5 dBm). Apply −50 dB=m, see 2.12 V (−13 dBm). We remove L1. We remove R20, R21, DA1, and DA2. We replace R9, R10, R12, R13 with 0 Ω. We apply − 60 dBm of 915 MHz and see −5 dBm on J4. We remove R19 and replace R6 and R7 with 0 Ω. Apply −50 dBm to J1 and see P = 2.16 V (−12 dBm). Restore R19, R6, R7 and see P = 2.08 V. Removing our 3-dB attenuator between L2 and U2 gives us a 2-dB increase in gain. Of our six attenuators, we leave the first three in place to stabilize the antenna and filter. We remove the fourth and fifth to increase the overall gain. We leave the sixth one in place because it makes no difference to the relative power of any signal, and offers better isolation of the switched capacitor filter. We apply −58 dBm SCT signal to J1 with L1 loaded. When interference dies down we have a clear 200 mVpp SCT signal on D. With −64 dBm we never get a clear signal.

[23-SEP-20] We measure the reflection coefficient of J1 with L1 loaded and R1 = R2 = 8.2 Ω, R17 = 150 Ω. We apply −20 dBm to the OUT terminal of a ZFDC-10-5+ directional coupler. We connect CPL to our spectrometer 50-Ω input. We connect IN to a 1-m coaxial cable. With 50-Ω termination we see −52 dBm at CPL, suggesting reflection of no more than −41 dBm, or 1% of incident power. We remove terminator and see −29 dBm, suggesting −18 dBm reflected. We call that 100%. now conect to J1 and see −48 dBm on CPL, suggesting −37 dBm reflected power, or roughly 2% reflection.

We remove L2, the SAW filter, so we can see the response of the demodulation tuner. We apply a −26 dBm sweep.


Figure: Demodulator (Green) and Detector (Blue) Output for −26 dBm Sweep Input, No SAW Filter (L2). We apply sweep to A, L1 loaded. Red trace is sweep mixed with 915 MHz and low-pass filtered to ±21 MHz.

We adjust C16 and C17. We have L3 = 3.3 nH. We measure the frequency and height of the peak in D


Figure: Demodulator Peak Frequency and Voltage for C16 and C17. No SAW filter (L2) loaded, apply −26 dBm sweep.

We restore C16 = 0.5 pF, C17 = 2.0 pF and restore L2, the SAW filter. We apply a −26 dBm sweep.


Figure: Demodulator (Green) and Detector (Blue) Output for −26 dBm Sweep Input, With SAW Filter (L2). We apply sweep to A, L1 loaded. Red trace is sweep mixed with 915 MHz and low-pass filtered to ±21 MHz.

[24-SEP-20] We apply 4.5 V power supply to our A3038X by mistake for twenty seconds, but the board seems fine afterwards. Current consumption 33 mA at 3.0 V supply. We replace L2, the SAW filter. We remove L1. We use J2 to bring B to a ZAD-11 RF input, and mix with 820 MHz LO. We choose our LO frequency so it is well outside the pass-band of the SAW filter. We obtain a clean sweep on D for 1000 MHz LO as well. We view the IF on the scope, as well as D and P.


Figure: Demodulator (Green) and Detector (Blue) Output for −60 dBm Sweep with New SAW Filter (L2). We apply sweep to A, L1 not loaded. Red trace is B mixed with 820 MHz.

We load L1 and connect a −3 dBm sweep to an A3015C loop antenna. We place the antenna 20 cm from L1. We wait for interference to die down and take the following picture.


Figure: Demodulator (Green) and Detector (Blue) Output for Sweep Picked Up By Antenna L1. We apply −3 dB sweep to an A3015C loop antenna standing 20 cm from L1. Red trace is B mixed with 820 MHz.

We remove L1 and apply −48 dBm SCT signal to J1. When interference subsides, we get clear SCT logic levels on D.